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? semiconductor components industries, llc, 2000 september, 2000 rev. 0 1 publication order number: and8024/d and8024/d off-line critical conduction switching power supply with voltage and current limiting prepared by: larry hayes, jim spangler on semiconductor phoenix, az 85008 abstract the need for a small, economical solution for switching power supplies and for battery chargers has increased. these applications generally require both voltage limiting and current limiting. these must be very efficient and produce a low amount of emi radiated noise. the mc33364 is a critical conduction control ic that can be configured in a resonant turnoff mode to reduce the amount of emi generated during turnoff of the power mosfet. the mc33341 is a secondary side controller that contains the band gap reference to provide a very tight tolerance voltage reference. the mc33341 also allows for current limiting thus providing an almost square loop of voltage and current needed for lithiumion battery charging. combining these two devices produces a much more costeffective solution than other approaches offer. the classic approach in prior years, a very cost efficient and economical design approach was demanded for battery chargers and power supplies. the unit had to be current limited on the secondary side and provide the proper voltage output. the supply often did not support universal input. one circuit approach to meet this need is shown in figure 1. this is a classical blocking oscillator. this circuit has its roots back to the old vacuum tube days, and was updated by bob haver in 1984 and presented at various switching power supply seminars [1], [2]. this is a variable frequency unit that operates in what is known today as the critical conduction mode. the unit starts by having the gate of the tmos or power mosfet turned on by the series resistors r1 and r2. the current through the primary of the coupled inductor transformer ramps up linearly. the auxiliary winding on the transformer is phased to provide positive (in phase) gate voltage. the tmos device is aturned offo when the primary current produces approximately 0.7 volts across the source resistor (rsource), the bipolar junction transistor (bjt) becomes forward biased and aturns ono. when the bjt is aturned ono, the charge on the gate is removed and the tmos power fet is aturned offo. the auxiliary voltage is reversed as the voltage on the primary is reversed in order to release its stored energy to the secondary. as long as there is energy flowing to the secondary, the auxiliary voltage is negative and the gate voltage is negative. the small signal diode (1n914) in series with the collector of the bjt keeps the device from operating in a reverse mode. in lieu of 1n914, a 1n4148 or 1n4150 may be used. with the tmos power fet in the aoffo mode, the coupled energy stored in the air gap of the inductor is released to the secondary. the secondary energy flows to the load by way of schottky diode d2. when the coupled energy is depleted, the voltage across the both the primary, secondary and auxiliary windings decreases to zero. the transformer is not a perfect device, so there is a small amount of energy remaining in the primary. this is the energy in the leakage inductance of the primary. this causes the voltage across the transformer to start to ring with the c oss capacitance of the fet. this aturnoffo ring has the effect of reducing the drainsource voltage of the fet, and causes the auxiliary winding to develop a positive voltage. this positive voltage, along with two resistors r1 and r2 places a positive voltage on the gate of the tmos power fet; this restarts the energy storage cycle. http://onsemi.com application note
and8024/d http://onsemi.com 2 ?? ?? r1 r2 q2 a1 rs r4 r5 a2 r6 r7 boost supply +output r8 q1 snubber c1 mtp3n60e d2 z tl431 figure 1. blocking oscillator with voltage and current limit on the secondary side + + output rsource 1n914 4 8 ac input classic voltage and current feedback the secondary side contains all the control functions. the voltage feedback is determined by the classic tl431 with resistors r6 and r7. the voltage across r7 is 2.54 volts. the current through the led of the optocoupler causes the npn phototransistor to conduct, thereby developing a small voltage across the resistor r8, which produces an offset voltage to the base of q1. this causes the primary current to be reduced during a switching or energy storage cycle. the current limiting is accomplished by the section to the left of the voltage divider resistors r6 and r7. the series resistor, r s , has the same voltage across it as r4. the ratio of r4/r s can be thought as a current gain. the operational amplifier, a1, provides current to the base of a high h fe npn transistor to maintain vr4 equal to vr s . (in some applications q2 is an nchannel jfet.) the current through r4 is the same current through r5. when voltage across r5 (v r5 ) exceeds 2.54 volts, the buffer stage composed of a2 and the series diode increases the voltage across r7, which causes the led current to increase and raises the offset voltage on the primary side. only when v r5 exceeds the tl431 reference voltage is the current limit applied. in order to have the operational amplifiers a1 and a2 (mc33072) function properly, a boosted supply voltage is necessary. this is accomplished by using the forward voltage mode of the transformer. this boosted supply voltage is limited by the zener, z. the maximum load current can be defined by equation 1: i loadmax 2.5 r5 * r4 rs (1) as an example, if the maximum is 3.0 amps, r s is chosen to dissipate less than 0.5 watts at full load. this calculates to be a 0.05556 ohm resistor, therefore a more practical value of 0.05 ohm is chosen, which is typically composed of two 0.1 ohm resistors. the next item to be chosen is the current gain. a value of 200 would produce a maximum current of 15 ma through r4 and r5. this would make r4 equal to 10 ohms (0.05 * 200 = 10). the value of r5 then becomes 166.667 ohms. this value can be created by using two series resistors, a 160 ohm and a 6.8 ohm, both of which are 0.25 watt or less. the effect of the feedback is to create a square output characteristic as shown in figure 2. because there is no perfect square output characteristic, the current is raised slightly. in this case only a 150 ohm resistor would be used for r5 which raises the output current to 3.33 amps. a 160 ohm resistor would provide 3.125 amps.
and8024/d http://onsemi.com 3 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 0123456789 output voltage (v) load current (a) figure 2. square voltage and current output characteristics problems and issues with the blocking oscillator there are many issues associated with the blocking oscillator. the first issue is that it can only be designed for a single voltage input, such as 120 vac or 220 vac. this will not operate over the entire range from 85 vac to 270 vac input, due to the auxiliary voltage variation. the auxiliary voltage is proportional to the rectified dc line voltage across the input electrolytic capacitor c1. if the auxiliary voltage is 7 volts with the valley of voltage of c1 equal to 100 volts, this would mean 13 volts for 132 vac input and 21.77 volts for a 220 vac input. there are two problems with this. first, 7 volts is very near the threshold of the high voltage power fet for good saturation. the lowest voltage for a nonlogic level device using aengineering rule of thumbo is 8 volts. the highest gate voltage which should be applied using good engineering practice is 8 volts. clearly, this is not a auniversal inputo candidate. the second issue is the gate turnoff. the transistor q1 has a limit to its ability to rapidly aturnoffo the power fet. the power fet therefore dissipates additional energy, requiring a large heat sink. the third issue occurs under a no load condition, where the frequency of operation becomes very high. problems and issues with the classic secondary side control the initial problem is operation under a short circuit condition. when the led is connected to the +output, there is no voltage to power the led to cut back on the short circuit current. therefore, the led should be connected to the aboosted supplyo. the current should be limited through the led to maintain a voltage on the aboosted supplyo. the second issue is the large number of components necessary to implement this circuit. even though all of the components are available as surface mount components, it is important to note that quality, cost and space are usually impacted as component count increases. the improved approach primary side integrated circuit, mc33364 today, a power supply is required have a universal input. laptop computer, cellular phone, and various other pieces of equipment are required to be able to operate in any country or place on earth where ac or dc electrical energy is available. the requirement today is to only change the plug, which connects to power supply. the requirements are: 1. universal input voltage 2. circuit short operation 3. open circuit operation 4. power limit or square loop load operation there are various integrated circuits today that can perform this function; one of them is the mc33364. the mc33364 is labeled a critical conduction controller, which is the same as the traditional blocking oscillator shown in figure 1. therefore, the transformer design is identical to that of the blocking oscillator. the advantages of this ic over the discrete blocking oscillator are: 1. wide input voltage range from 40 vdc to 700 vdc. this ic can operate from the 48 volt telecom dc bus to the 381 vac linetoline voltage common in europe 2. full gate drive to properly drive high voltage power fet or igbts in a to220 package that only need 15 volts 3. frequency clamp to limit the open circuit frequency 4. reference to be able to perform primary side control
and8024/d http://onsemi.com 4 the essential difference between the traditional and the mc33364based circuits is the phasing of the auxiliary winding. the auxiliary winding is phased the same as the output winding. this allows a control auxiliary supply voltage that can be tightly coupled to the primary and secondary. secondary controller ic, mc33341 the mc33341 is an 8 pin device the same size as the dual operational amplifier mc33072 in the discrete, classic system controller. an internal reference is set at approximately 1.2 volts which allows for low voltage applications. this is similar to the tlv431. the advantages of the mc33341 approach are as follows: 1. small size 2. fewer components 3. led is driven directly from the ic for easier short circuit operation 4. positive (+ bus) or negative ( bus) sensing there are the following similarities: 1. resistor divider network for voltage determination 2. two resistors for current sensing 3. need for a boosted supply for short circuit operation design example: an 85 to 270 vac in; 3 a, 8.2 v out controller a complete design is shown in figure 3. this implementation has the same function as that shown in figure 1, except that now universal input is supported. this supply will operate over the input range from 85 vac to 270 vac. the resultant power supply or battery charger will have a square loop output characteristic with the ability to supply 3.0 amps continuously at 8.2 volts. a full load frequency of 70 khz at low line is chosen. the output is: watts load 3.0 * 8.2 24.6 watts watts input watts load (efficiency) 24.6 0.85 28.9 watts (2) (3) for this design, we will use 30 watts to keep the numbers simple. 12 transformer boosted supply +output output snubber mc33364 mc33341 4n35 mmbd914 mur180 47 k 8 1 45 22 murs120 1 k 5.1 zener .01 mbrs360 150 8.2 v figure 3. universal input controller with an output at 3 amps and 8.2 volts mtp3n60e ac input 100 f 10 f 1 rsource .01 f 1.0 f 2000 f 47 f 0.47 f 16 9 10 11 6 8 4 3
and8024/d http://onsemi.com 5 low line is an 85 vac, 50 hz input with a 20 volt ripple on the electrolytic at this voltage. this defines the valley, from which the size of the electrolytic capacitor and the value of the inductor are determined. the value of the capacitor can be determined by several methods. two will be given here. the first comes from a text by savant et al [4] where: c v peak at lowline v peak ripple *f re q *r l o ad (4) 85 * 2 20 * 100 * 500 120 f for a 20 volt peaktopeak ripple (5) 85 * 2 25 * 100 * 500 96 f for a 25 volt peaktopeak ripple (6) r load v 2 cap average watts (7) v cap average v peak at lowline v peak ripple 2 (8) the second method uses the joules of energy to calculate the value of the capacitor. from engineering experience, the length of time the diodes conduct in a full wave bridge is approximately 2.5 milliseconds for a 50 hz system and about 2.25 milliseconds for a 60 hz system. this means the energy storage electrolytic capacitor must supply the energy during the nonconducting period. this is 7.5 milliseconds for 50 hz and 6 milliseconds for a 60 hz system. c 2 * 0.225 85 * 2 2 85 * 2 25 2 83.5 f c 2 * joulesload v 2 peak v 2 valley joules 1 2 cv 2 peak 1 2 cv 2 valley joules 30 * 7.5e 3 0.225 joules watt * seconds (9) (10) (11) (12) (13) both methods indicate a large electrolytic capacitance value is needed for operation at 85 vac 50 hz. the dc rating of the electrolytic capacitor is defined by the ac high line voltage, which is 270 volts. this means a 400 volt electrolytic capacitor is necessary. the typical embodiment is one 100 m f 400 vdc electrolytic, or two 47 m f 400 vdc electrolytics in parallel. as an added note, if the supply were to be used in the us and canada on a 120 volt, 60 hz line, 2 times load is an aengineering rule of thumbo. this would be a 60 m f capacitor, or a 68 m f from manufacturers' catalogs. if in europe, with its 220 vac, 50 hz line, the capacitor would be the 30 m f, or as a typical catalog value, 33 m f. power factor correction (pfc) and hold up time the design of a universal input from 85 vac 50 hz to 270 vac 50 hz can be dif ficult, as seen by the size of electrolytic capacitors. the above capacitor sizing does not include any oholdupo time for missing ac line cycles. if a oholdupo time is required along with a universal input, it is strongly suggested that an active pfc circuit be considered. this would reduce the capacitance value of the electrolytic capacitor and reduce the current rating of the flyback power mosfet. the voltage ratings of both parts would remain nearly identical. using an active pfc circuit like the mc33262 or the mc33261 could help reduce the physical size of the unit. the data sheets for the mc33262 and mc33261 contain design equations, and several reference designs are included with data. transformer or coupled inductor design because the minimum voltage is defined, the value of the inductance can now be calculated. the minimum dc voltage across the electrolytic capacitor is 95 volts. at this point, the duty cycle is a maximum of 0.5, meaning that the ontime is equal to the offtime of the switch. the peak current for the switch and the inductance value are both calculated. this is developed by chryssis [5]. the minimum frequency at full load is calculated to be 70 khz. i peak primary 2 * watts v dc.min * max (14) 2*30 95 * 0.5 1.26 amps (15) e l i t l 2*watts v dc.min * max freq (16) l v 2 dc.min * 2 max 2 * watt * freq 95 2 *0.5 2 2*30*70e3 537 h (17) the core was chosen from parts readily available; ferrite international (7070301507), magnetic (p43007), philips (e30/15/73c80). this core has the following parameters: cross sectional area, ae or ac = 0.49 cm 2 min. center leg magnetic material path length, lm = 6.56 cm
and8024/d http://onsemi.com 6 the turns are calculated using equation 19, which is developed from equation 18. e n t na c b t (18) n e t a c b (19) for the primary, there are 68 turns using awg 25 wire. next, the air gap is calculated to determine the edge space. the gap is calculated using equations 20 and 21. the gap is the total gap including the edge gap and the center gap. since ungapped cores are being used, the gap length is divided by two, and the spacer is placed on the outer legs. the total gap is 2.05 e5 meters, or 0.008 inches. this means a spacer of 0.004 inches, which is the thickness of sheet of copier paper. l n 2 n 2 core material n 2 l m 0 r a c l gap 0 a gap (20) l gap 0 a c n 2 l l m r (21) l = inductance in henries l m = core magnetic path length in meters l gap = gap length in meters a c =a e =effective cross sectional area in meters 2 u o = 4 10 7 u r = core permeability n = number of turns the primary voltage divided by the primary turns defines the volts per turn, which is approximately 1.4 volts per turn. the secondary turns is calculated to be 6.35 turns. a value of 7 turns is chosen for the 8.3 volt secondary using two strands of 24 awg magnet wire, and 10 turns is chosen for the auxiliary winding, using 30 awg. if the power switch is an igbt requiring 15 volts, the auxiliary winding would be 13 turns. all of this easily fits onto the single section bobbin. additional information is available in the form of application notes from on semiconductor, see references [6], [7], and [8]. transformer sizing it is beyond the intent of this application note to teach exact transformer and inductor design. the information given is used as a guide for background information. choosing the size of the transformer or coupled inductor is easy, but there are rules of thumb. one of the easier methods is to use the oarea producto method. the equations are developed in the magnetics ferrite core catalog [9]. another set of equations along with data is found in c. mclyman's books [10], [11], [12], which have become standards for transformer and inductor designers. the area product (a p ) equation is defined below for single transistor forward converter transformers and flyback transformers. this defines the smallest possible core. a p = area product in cm 4 p o = output power of all the windings c = inverse of current density 1/j = 3.55*10 3 cm 2 /amp a p p o c*10 8 2 e bfk (22) this is similar to 500 circular mils per amp. e = efficiency, often taken as 0.9 for a starting point b = peak flux density the core will see f = switching frequency k = transformer utilization factor, typically 0.3 for a transformer this can never exceed 0.5 for an inductor this can be near 0.8 a p 3*3.22*10 3 *10 8 2* 0 . 9 *2 000 *7 0 *1 0 3 * 0 . 3 0.12778cm 4 (23) the core chosen must be larger than 0.127 cm 4 . in order to meet many of the european regulator standards, the a p may have to be increased by a factor between 2 and 3. this is where the art of transformer design begins to play an important part. the core chosen for this example has an a p = 0.55 cm 4 .
and8024/d http://onsemi.com 7 nondissipated snubber design for quasi resonant mode operation the snubber circuit can be modified to make the circuit dissipate less energy and to lower the switching transitions. this is shown in figure 4. the lossless snubber described in chapter 11 of the rectifier applications handbook [13] reduces the turnoff losses of the power fet by limiting the dv/dt rise of the drainsource voltage. by starting with equation 24 and rearranging into equation 25, the turnoff energy to the power fet can be greatly reduced. nc 16 12 boost supply +output output mc33364 mc33341 4n35 1n4150 8 1 45 22 murs120 5.1 v mbrs360 150 figure 4. lossless snubber circuit mtp3n60 47 f 22 k rsource 1.0 f 1000 pf 1000 f 1000 f 10 f r6 r7 0.01 m f 0.05 w nc nc 10 9 1 nc 3 4 8 6 11 mur120 47 lossless snubber lr 1000 pf cr 470 1n4006 1n4006 1n4006 1n4006 10 f ds2 mur180 ds1 mur180 1.25 v 470 during the beginning of the power fet aturnoffo period, the primary current wants to continue to flow, and forces the primary winding to change voltage direction. instead of the primary current flowing through the fet, the current is directed through the capacitor, cr, and diode, ds1. the capacitor, cr, charges to a voltage defined by equation 26. the energy stored is given by equation 27. i c v t (24) (25) i c c v t (26) v cr ( v output 0.7 v ) n primary n secondary 86.4 (8.2 0.7) 68 7 (27) j cr 1 2 cv 2 cr after all of the coupled energy is released to the secondary, the voltage on the auxiliary winding rings through zero and forces the mc33364 to start the next energy storage cycle. when the power fet turns on, the drain of the fet is at ground potential. there is a complete circuit consisting of the fet, diode (ds2), inductor (lr), and capacitor (cr). in an ideal condition the voltage on the capacitor reverses its polarity, this is explained by rashid in chapter 2 of his text opower electronicso [14]. the time for this transition to occur is defined by equation 28. (28) t cr * lr the value of the peak current is defined by equation 29. (29) i r v cr cr lr there are several limits that need to be considered. the time needs to be less than 1 microsecond. the value of the peak current needs to be less than the maximum peak current the power fet can handle. table 1 shows this variation where cr is 1000 picofarads and v cr = 86.4 volts using equation 26. the construction of the inductor lr is somewhat critical. the inductor should be a toroid using a powered iron material with a u r less than 100. the reason for a toroid is to
and8024/d http://onsemi.com 8 confine the flux in small space and not radiate to the rest of the circuit. using a micrometals [15] toroidal core, t3826 or t3852, with 14 turns, an 9.6 microhenry inductor can be constructed. this corresponds to a transition time of just over 0.3 microseconds and a peak current of 0.9 amps. this is considerably under the peak current rating of the mtp3n60e device. table 1. lr and i peak for given times time in s lr in h i peak 0.2 4.05e 06 1.357 0.3 9.12e 06 0.905 0.4 1.62e 05 0.679 0.5 2.53e 05 0.543 0.6 3.65e 05 0.452 0.7 4.96e 05 0.388 0.8 6.48e 05 0.339 0.9 8.21e 05 0.302 1.0 1.01e 04 0.271 there is an rc filter between the fet source sense resistor, rsource, and the input to the ic, mc33364. the time constant of this rc network needs to be greater than the transition time of the snubber network. the purpose of this rc filter is to eliminate false triggering, and is often called leading edge blanking network.
and8024/d http://onsemi.com 9 references [1] bob haver, semiconductors for switching power supplies, seminar handout 1985, ? 1985, motorola inc. [2] jim spangler, aa power factor corrected mosfet, multiple output, flyback switching supplyo, proceedings 10th pcim '85 conf. oct 85, pp 1932. [3] data sheets, mc33364, mc33341, tl431, mc33072, mtp1n60e, mmbt2222, mbrs360, mra4004, murs160, mbt4n60e, ntd3n60 from on semiconductor, http://www.onsemi.com [4] savant, roden, carpenter, aelectronic design, circuits and systems, 2nd ed o, ? 1987, 1991, benjamin/cummings publishing co., redwood city, ca, isbn 0805302859, pp 3842. [5] chryssis, george c., ahighfrequency switching power supplies, design & theory o, ? 1989, 1994, mcgrawhill, ny, isbn 0070109516, chapter 5. [6] ondrej pauk, an1593, alow cost 1.0 a current source for battery chargerso, ? 1997, motorola inc., http://www.onsemi.com [7] c. basso, an1681, ahow to keep a flyback switch mode supply stable with a criticalmode controllero, ? 1999, motorola inc., http://www.onsemi.com [8] petr lidak, unpublished application note in 1996 aacdc battery charger (constant current with voltage limit) using the mc33364 and the mc33341. [9] magnetic ferrite cores catalog, butler, pa, 7242826955. http://maginc.com [10] c. mclyman,atransformer and inductor design handbooko, ? 1978, marcel dekker, new york, ny; isbn 0824768019 [11] c. mclyman, amagnetic core selection for transformers and inductorso, ? 1982, marcel dekker, new york, ny; isbn 0824718739 [12] c. mclyman, ? 1987, designing magnetic components for high frequency dcdc converters, kg magnetics, inc., san marino, ca; isbn 1883107008 [13] rectifier applications handbook, hb214/d, ? 1993, isbn 0929270002; available from on semiconductor, phoenix, az 85008 [14] rashid, apower electronics, circuits, devices, and applicationso; ? 1988, prentice hall, englewood cliffs, nj 07632, isbn 0136876676, pp.1823 [15] micrometals iron powered cores, catalog 4 issue g, ? 1993, anaheim, ca, 8003565977
and8024/d http://onsemi.com 10 notes
and8024/d http://onsemi.com 11 notes
and8024/d http://onsemi.com 12 on semiconductor and are trademarks of semiconductor components industries, llc (scillc). scillc reserves the right to make changes without further notice to any products herein. scillc makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does scillc assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. atypicalo parameters which may be provided in scill c data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. all operating parameters, including atypicalso must be validated for each customer application by customer's technical experts. scillc does not convey any license under its patent rights nor the rights of others. scillc products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body , or other applications intended to support or sustain life, or for any other application in which the failure of the scillc product could create a sit uation where personal injury or death may occur. should buyer purchase or use scillc products for any such unintended or unauthorized application, buyer shall indemnify and hold scillc and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthori zed use, even if such claim alleges that scillc was negligent regarding the design or manufacture of the part. scillc is an equal opportunity/affirmative action employer. publication ordering information central/south america: spanish phone : 3033087143 (monfri 8:00am to 5:00pm mst) email : onlitspanish@hibbertco.com asia/pacific : ldc for on semiconductor asia support phone : 3036752121 (tuefri 9:00am to 1:00pm, hong kong time) toll free from hong kong & singapore: 00180044223781 email : onlitasia@hibbertco.com japan : on semiconductor, japan customer focus center 4321 nishigotanda, shinagawaku, tokyo, japan 1410031 phone : 81357402745 email : r14525@onsemi.com on semiconductor website : http://onsemi.com for additional information, please contact your local sales representative. and8024/d north america literature fulfillment : literature distribution center for on semiconductor p.o. box 5163, denver, colorado 80217 usa phone : 3036752175 or 8003443860 toll free usa/canada fax : 3036752176 or 8003443867 toll free usa/canada email : onlit@hibbertco.com fax response line: 3036752167 or 8003443810 toll free usa/canada n. american technical support : 8002829855 toll free usa/canada europe: ldc for on semiconductor european support german phone : (+1) 3033087140 (monfri 2:30pm to 7:00pm cet) email : onlitgerman@hibbertco.com french phone : (+1) 3033087141 (monfri 2:00pm to 7:00pm cet) email : onlitfrench@hibbertco.com english phone : (+1) 3033087142 (monfri 12:00pm to 5:00pm gmt) email : onlit@hibbertco.com european tollfree access*: 0080044223781 *available from germany, france, italy, england, ireland tmos is a trademark of semiconductor components industries, llc (scillc).

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